1. Field of the Invention
The present invention relates to digital communications, and more particularly to a channel memory length selection method for wireless communication systems.
2. Description of the Prior Art
Transmission channels suffer from time-varying frequency selective fading in many wireless communication environments. The time-varying frequency selective fading in turn causes inter-symbol interference (ISI) problem at the receiving side of a wireless communication system. Before the transmitted data sequence being estimated in an equalizer, the channel memory length as well as the impulse response of current transmission channel should be estimated as accurate as possible.
In practice, wireless communication systems may operate in channel profiles having very different delay spread. In GSM/EDGE (Global System for Mobile communication/Enhanced Data rates for Global Evolution), for example, the largest delay may be 0.5 us (micro-second) in a rural area (RA) model, but may be up to 20 us in a hilly terrain (HT) model. Since the GSM symbol duration is about 3.69 us, the largest delays of the two models will last to about first and sixth taps of symbols respectively. This phenomenon makes it difficult to settle a constant channel memory length for all potential channel profiles. Both over-estimation and under-estimation of the channel memory length will lead to a degradation of the equalizer.
FIG.1 shows a schematic block diagram of a typical equalizer 100 in accordance with prior arts. Equalizer 100 contains a channel estimation unit 110 and a data estimation unit 120. The channel estimation unit 110 takes a received signal r(k) and a training sequence b(k) as inputs and outputs an estimated channel impulse response (CIR) h(k) to the data estimation unit 120 for subsequent equalization processing. In GSM protocol, for example, each burst of received signal r(k) in average contains 156.25 bits in the corresponding time slot. The training sequence b(k) is a known pattern resided in a burst for reconstruction of transmitted signals. Those skilled in the art will appreciate that the taps of CIR estimation h(k) may be obtained from the cross-correlation of the received signal r(k) and the training sequence b(k) as shown in the formula below:
            h      ⁡              (        k        )              =                  1        N            ⁢                        ∑                      i            =            0                                N            -            1                          ⁢                                  ⁢                              b            ⁡                          (              i              )                                *                      r            ⁡                          (                              k                +                i                            )                                            ,          ⁢      k    =    0    ,  1  ,            ⋯      ⁢                          ⁢      L        -    1  in which N is a properly selected number according to system design and L is referred to a number of the CIR taps capable of covering the worst case communication environment. For example, N may be 26 to use all the training sequence bits to estimate the CIR. Usually, a number less than 26 may be used under efficiency consideration. Other methods well known to those skilled in the art can also be used to estimate the CIR.
FIG.2 shows a known method to select an appropriate channel memory length best fitting the actual communication environment. The method includes computing an initial CIR estimation (step 20); determining a refined CIR of length LM with a maximum energy EM by sliding widow searching the initial CIR estimation (step 22); eliminating taps less than the product of a ratio R and the maximum energy EM in the refined CIR, in which the energy of a CIR may be evaluated by a sum of all taps of the CIR (step 24). The number of taps of the resulted refined CIR is selected as the channel memory length (step 26), and the refined CIR can be used for operation in subsequent stages to the equalizer.
The foregoing method does not consider the variation between different environments. In a typical low delay spread environment, most energy may concentrate in a limited number of taps, therefore a larger value of the ratio R may be required to remove redundant taps which are essentially introduced by fading or noise. On the contrary, in a high delay spread environment, energy may spread over more channel taps. In such case, a smaller ratio R may be required.